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LTC3561A 1A, 4MHz, Synchronous Step-Down DC/DC Converter FEATURES

DESCRIPTION
The LTC(R)3561A is a constant frequency, synchronous step-down DC/DC converter. Intended for medium power applications, it operates from a 2.5V to 5.5V input voltage range and has a user configurable operating frequency up to 4MHz, allowing the use of tiny, low cost capacitors and inductors 1mm or less in height. The output voltage is adjustable from 0.8V to 5.5V. Internal synchronous power switches provide high efficiency. The LTC3561A's current mode architecture and external compensation allow the transient response to be optimized over a wide range of loads and output capacitors. To further maximize battery life, the P-channel MOSFET is turned on continuously in dropout (100% duty cycle). In shutdown, the device draws <1A.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6580258, 6498466, 6611131.

Uses Tiny Capacitors and Inductor High Frequency Operation: Up to 4MHz Low RDS(ON) Internal Switches: 0.15 High Efficiency: Up to 96% Stable with Ceramic Capacitors Current Mode Operation for Excellent Line and Load Transient Response Short-Circuit Protected Low Dropout Operation: 100% Duty Cycle Low Shutdown Current: IQ 1A Low Quiescent Current: 330A Output Voltages from 0.8V to 5V VIN: 2.5V to 5.5V Small 8-Lead 3mm x 3mm DFN Package
APPLICATIONS

Notebook Computers Digital Cameras Cellular Phones Handheld Instruments Board Mounted Power Supplies
TYPICAL APPLICATION
Step-Down 2.5V/1A Regulator
VIN 2.5V TO 5.5V 10F PVIN SVIN SW LTC3561A ITH 16.9k 680pF 549k SHDN/RT SGND PGND 118k VFB 22pF 2.2H VOUT 2.5V 1A 22F EFFICIENCY (%)
Efficiency and Power Loss vs Output Current
100 90 80 70 60 50 40 30 20 10 0 1 VOUT = 2.5V fO = 1MHz VIN = 2.7V VIN = 3.6V VIN = 4.2V 100 1000 10 OUTPUT CURRENT (mA) 0.01 0.1 POWER LOSS (W) 1
249k
0.001
0.0001 10000
3561a TA01a
3561A TA01b
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LTC3561A ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW SHDN/RT 1 SGND 2 SW 3 PGND 4 9 8 ITH 7 VFB 6 SVIN 5 PVIN
PVIN, SVIN Voltages ..................................... -0.3V to 6V VFB, ITH, SHDN/RT Voltages ..........-0.3V to (VIN + 0.3V) SW Voltage ..................................-0.3V to (VIN + 0.3V) Operating Junction Temperature Range (Notes 2, 5, 8) ........................................ -40C to 125C Storage Temperature Range................... -65C to 125C Lead Temperature (Soldering, 10 sec) .................. 300C
DD PACKAGE 8-LEAD (3mm 3mm) PLASTIC DFN TJMAX = 125C, JA = 43C/W, JC = 3C/W EXPOSED PAD (PIN 9) IS SGND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3561AEDD#PBF LTC3561AIDD#PBF TAPE AND REEL LTC3561AEDD#TRPBF LTC3561AIDD#TRPBF PART MARKING* LDKB LDKB PACKAGE DESCRIPTION 8-Lead (3mm x 3mm) Plastic DFN 8-Lead (3mm x 3mm) Plastic DFN TEMPERATURE RANGE -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V, RT = 125k unless otherwise specified. (Note 2)
SYMBOL VIN IFB VFB VLINEREG VLOADREG gm(EA) PARAMETER Operating Voltage Range Feedback Pin Input Current Feedback Voltage Reference Voltage Line Regulation Output Voltage Load Regulation Error Amplifier Transconductance (Note 3) (Note 3) VIN = 2.5V to 5.5V ITH = 0.55V to 0.9V ITH Pin Load = 5A (Note 3)

ELECTRICAL CHARACTERISTICS
CONDITIONS
MIN 2.5 0.784
TYP
MAX 5.5 0.1
UNITS V A V %/V % S
0.8 0.04 0.02 300
0.816 0.2 0.2
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LTC3561A ELECTRICAL CHARACTERISTICS
SYMBOL IS VSHDN/RT fOSC ILIM RDS(ON) ISW(LKG) VUVLO tSOFT-START PARAMETER Input DC Supply Current (Note 4) Active Mode Shutdown Shutdown Threshold High Active Oscillator Resistor Oscillator Frequency Peak Switch Current Limit Top Switch On-Resistance Bottom Switch On-Resistance Switch Leakage Current Undervoltage Lockout Threshold
The denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 3.6V, RT = 125k unless otherwise specified. (Note 2)
CONDITIONS VFB = 0.75V VSHDN/RT = 3.6V MIN TYP 330 0.1 VIN - 0.6 125k 2.5 2.0 0.15 0.13 0.01 1.8 0.5 2.1 0.8 MAX 450 1 VIN - 0.4 1M 2.8 4 2.5 0.18 0.16 1 2.4 1 UNITS A A V MHz MHz A A V ms
RT = 125k (Note 7) VFB = 0.5V (Note 6) (Note 6) VIN = 5V, VSHDN/RT = 3.6V, VSW = 0V or 5V VIN Ramping Down 10% to 90% of Regulation
2.25 1.3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3561AEDD is guaranteed to meet specified performance specifications from 0C to 85C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3561AIDD is guaranteed over the full -40C to 125C operating junction temperature range. Note 3: The LTC3561A is tested in a feedback loop which servos VFB to the midpoint for the error amplifier (VITH = 0.7V). Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency.
Note 5: TJ is calculated from the ambient TA and power dissipation PD according to the following formulas: TJ = TA + (PD * 43C/W) Note 6: Switch on-resistance is sampled at wafer level measurements and assured by design, characterization and correlation with statistical process controls. Note 7: 4MHz operation is guaranteed by design but not production tested and is subject to duty cycle limitations (see Applications Information). Note 8: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability.
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LTC3561A TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25C, VIN = 3.6V, fO = 1MHz, unless otherwise noted. Efficiency vs Input Voltage
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 2.5 3.0 4.0 4.5 3.5 INPUT VOLTAGE (V) 5.0 5.5 IOUT = 10mA IOUT = 1A EFFICIENCY (%) IOUT = 100mA VOUT = 1.8V 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 100 1000 10 OUTPUT CURRENT (mA) 10000
3561A G02
Efficiency vs Output Current
VOUT = 1.8V 100 90 80 70 60 50 40 30 20 10 0
Efficiency vs Output Current
VOUT = 1.5V
VIN = 2.7V VIN = 3.6V VIN = 4.2V 1 100 1000 10 OUTPUT CURRENT (mA) 10000
3561A G03
3561A G01
Efficiency vs Frequency
95 94 93 EFFICIENCY (%) 2.2H 92 91 1H 90 89 88 0 1 3 2 FREQUENCY (MHz) 4.7H VOUT ERROR (%) 0.2 0.1 0 -0.1 -0.2 4 5
3561A G05
Load Regulation
0.4 0.3 VOUT ERROR (%) VOUT = 1.8V 0.6 0.4 0.2 0.0 -0.2 -0.4
Line Regulation
VOUT = 1.8V ILOAD = 400mA
VOUT = 1.8V ILOAD = 400mA
0
200
400 600 800 1000 1200 1400 OUTPUT CURRENT(mA)
3561A G06
-0.6 2.5
3.0
4.5 4.0 3.5 INPUT VOLTAGE(V)
5.0
5.5
3561A G07
Reference Voltage vs Temperature
815 810 805 800 795 790 785 -50 6 4 2 0 -2 -4
Frequency Variation vs Temperature
6 4 FREQUENCY VARIATION (%) 2 0 -2 -4 -6
Frequency Variation vs VIN
FREQUENCY VARIATION (%)
REFERENCE VOLTAGE (mV)
-25
50 25 75 0 TEMPERATURE(C)
100
125
-6 -50
-25
50 25 75 0 TEMPERATURE(C)
100
125
-8 2.5
3.0
3.5
4.0 VIN (V)
4.5
5.0
5.5
3561A G10
3561A G08
3561A G09
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LTC3561A TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25C, VIN = 3.6V, fO = 1MHz, unless otherwise noted. RDS(ON) vs Input Voltage
0.25 0.30 0.25 MAIN SWITCH RDS(ON) () 0.15 SYNCHRONOUS SWITCH RDS(ON) () 0.20 MAIN SWITCH 0.15 SYNCHRONOUS SWITCH 0.10 0.05 0.05 0.0 -50 DYNAMIC SUPPLY CURRENT (A)
RDS(ON) vs Temperature
300
Dynamic Supply Current vs Input Voltage
VOUT = 1.8V 290 ILOAD = 0A 280 270 260 250 240 230 220 2.5 3 3.5 4 VIN (V) 4.5 5 5.5
3561A G13
0.20
0.10
0.0 2.5
3.0
4.5 4.0 3.5 INPUT VOLTAGE (V)
5.0
5.5
3561A G11
-25
50 25 75 0 TEMPERATURE (C)
100
125
3561A G12
Supply Current vs Temperature
360 340 SUPPLY CURRENT (mA) 320 300 280 260 240 220 200 -50 -25 0 75 50 25 TEMPERATURE (C) 100 125 VOUT = 1.8V ILOAD = 0A SWITCH LEAKAGE (pA) 2500
Switch Leakage vs Input Voltage
600 500 MAIN SWITCH 1500 SWITCH LEAKAGE (nA) 400 300
Switch Leakage vs Temperature
2000
1000
SYNCHRONOUS SWITCH 200 100 0 -50 MAIN SWITCH
500
SYNCHRONOUS SWITCH
0
0
1
4 3 2 INPUT VOLTAGE(V)
5
6
3561A G15
-25
50 25 75 0 TEMPERATURE (C)
100
125
3561A G14
3561A G16
Switching Waveforms
SHDN/RT 2V/DIV
Start-Up from Shutdown
SW 2V/DIV VOUT 50mV/DIV AC COUPLED IL 200mA/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 5mA 4s/DIV
3561A G18
VOUT 1V/DIV
IL 1A/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A 200s/DIV
3561A G20
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LTC3561A TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25C, VIN = 3.6V, fO = 1MHz, unless otherwise noted. Start-Up from Shutdown
SHDN/RT 2V/DIV VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 1A 200s/DIV
3561A G21
Load Step
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 0A to 1A
3561A G23
Load Step
VOUT 1V/DIV
IL 1A/DIV
40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 20mA to 1A
3561A G24
Load Step
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 200mA to 1A
3561A G25
VOUT Short to Ground
VOUT 1V/DIV
IL 2A/DIV
VIN = 3.6V VOUT = 1.8V ILOAD = 0A
40s/DIV
3561A G26
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LTC3561A PIN FUNCTIONS
SHDN/RT (Pin 1): Combination Shutdown and Timing Resistor Pin. The oscillator frequency is programmed by connecting a resistor from this pin to ground. Forcing this pin to SVIN causes the device to be shut down. In shutdown all functions are disabled. SGND (Pin 2): Signal Ground. All SGND and PGND pins must be connected together through a thick copper trace or ground plane. SW (Pin 3): The Switch Node Connection to the Inductor. This pin swings from PVIN to PGND. PGND (Pin 4): Power Ground. Connect to the (-) terminal of COUT, and (-) terminal of CIN. All SGND and PGND pins must be connected together through a thick copper trace or ground plane. PVIN (Pin 5): Main Supply Pin. Must be closely decoupled to PGND.
PIN 1 2 3 4 5 6 7 8 NAME SHDN/RT SGND SW PGND PVIN SVIN VFB ITH DESCRIPTION Shutdown/Timing Resistor Signal Ground Switch Node Main Power Ground Main Power Supply Signal Power Supply Output Feedback Pin Error Amplifier Compensation and Run Pin -0.3 2.5 0 0.4 0.8 0 0 5.5 5.5 1.0 1.4 -0.3 -0.3 -0.3 -0.3 SVIN + 0.3 6 SVIN + 0.3 SVIN + 0.3 MIN -0.3
SVIN (Pin 6): The Signal Power Pin. All active circuitry is powered from this pin. Must be closely decoupled to SGND. SVIN must be greater than or equal to PVIN. VFB (Pin 7): Receives the feedback voltage from the external resistive divider across the output. Nominal voltage for this pin is 0.8V. ITH (Pin 8): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0.4V to 1.4V. Exposed Pad (Pin 9): Signal Ground. All small-signal components and compensation components should be connected to this ground (see Board Layout Considerations). Must be soldered to electrical ground on PCB. All SGND and PGND pins must be connected together through a thick copper trace or ground plane.
NOMINAL (V) TYP 0.8 0
ABSOLUTE MAX (V) MAX SVIN PVIN MIN -0.3 -0.3 MAX SVIN + 0.3 PVIN + 0.3
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LTC3561A BLOCK DIAGRAM
SVIN 6 SGND 2 ITH 8 PVIN 5
0.8V
VOLTAGE REFERENCE
ITH LIMIT BCLAMP
PMOS CURRENT COMPARATOR
+ -
+ -
VFB 7
-
ERROR AMPLIFIER VB
+
BURST COMPARATOR SLOPE COMPENSATION 3 SW
OSCILLATOR
+
LOGIC
NMOS COMPARATOR
- - +
4 PGND
REVERSE COMPARATOR 1 SHDN/RT
3561A BD
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LTC3561A OPERATION
The LTC3561A uses a constant frequency, current mode architecture. The operating frequency is determined by the value of the RT resistor. The output voltage is set by an external divider returned to the VFB pin. An error amplifier compares the divided output voltage with the reference voltage of 0.8V and adjusts the peak inductor current accordingly. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle. Current flows through this switch into the inductor and the load, increasing until the peak inductor current reaches the limit set by the voltage on the ITH pin. Then the top switch is turned off, the bottom switch is turned on, and the energy stored in the inductor forces the current to flow through the bottom switch, and the inductor, out into the load until the next clock cycle. The peak inductor current is controlled by the voltage on the ITH pin, which is the output of the error amplifier. The output is developed by the error amplifier comparing the feedback voltage, VFB, to the 0.8V reference voltage. When the load current increases, the output voltage and VFB decrease slightly. This decrease in VFB causes the error amplifier to increase the ITH voltage until the average inductor current matches the new load current. The main control loop is shut down by pulling the SHDN/RT pin to SVIN, resetting the internal soft-start. Re-enabling the main control loop by releasing the SHDN/RT pin activates the internal soft-start, which slowly ramps the output voltage over approximately 0.8ms until it reaches regulation. Dropout Operation When the input supply voltage decreases toward the output voltage, the duty cycle increases to 100% which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drop across the internal P-channel MOSFET and the inductor. Low Supply Operation The LTC3561A incorporates an undervoltage lockout circuit which shuts down the part when the input voltage drops below about 2.1V to prevent unstable operation.
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LTC3561A APPLICATIONS INFORMATION
A general LTC3561A application circuit is shown in Figure 4. External component selection is driven by the load requirement, and begins with the selection of the inductor L1. Once L1 is chosen, CIN and COUT can be selected. Operating Frequency Selection of the operating frequency is a trade-off between efficiency and component size. High frequency operation allows the use of smaller inductor and capacitor values. Operation at lower frequencies improves efficiency by reducing internal gate charge losses but requires larger inductance values and/or capacitance to maintain low output ripple voltage. The operating frequency, fO, of the LTC3561A is determined by an external resistor that is connected between the RT pin and ground. The value of the resistor sets the ramp current that is used to charge and discharge an internal timing capacitor within the oscillator and can be calculated by using the following equation: RT 5 x 107 (fO)-1.6508 (k) where fO is in kHz, or can be selected using Figure 1. The maximum usable operating frequency is limited by the minimum on-time and the duty cycle. This can be calculated as: V fO(MAX) 6.67 * OUT (MHz) VIN(MAX) The minimum frequency is internally set at around 200kHz
5000 4500 4000 FREQUENCY (kHz) 3500 3000 2500 2000 1500 1000 500 0 0 400 800 1200 RT (k) 1600
3561A F01
Inductor Selection The operating frequency, fO, has a direct effect on the inductor value, which in turn influences the inductor ripple current, IL: IL = VOUT fO * L V * 1- OUT V IN
The inductor ripple current decreases with larger inductance or frequency, and increases with higher VIN or VOUT. Accepting larger values of IL allows the use of lower inductances, but results in higher output ripple voltage, greater core loss and lower output capability. A reasonable starting point for setting ripple current is IL = 0.4 * IOUT(MAX), where IOUT(MAX) is 1A. The largest ripple current IL occurs at the maximum input voltage. To guarantee that the ripple current stays below a specified maximum, the inductor value should be chosen according to the following equation: L= VOUT fO * IL VOUT * 1- V IN(MAX)
Inductor Core Selection Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don't radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price vs
TA = 25C
Figure 1. Frequency vs RT
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LTC3561A APPLICATIONS INFORMATION
size requirements and any radiated field/EMI requirements than on what the LTC3561A requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3561A applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER Toko A914BYW-1R2M=P3: D52LC A960AW-1R2M=P3: D518LC MAX DC VALUE CURRENT 1.2H 1.2H 2.15A 1.8A 2.1A 2.1A 2.1A 2.05A 3.3A 2.1A 2.3A 2.4A 2.1A 2.2A 2.2A 2.1A 3.0A 3.8A 2.56A 2.80A 2.04A 2.16A 2.6A 3.2A 2.9A 3.2A 1.4A DCR 44m HEIGHT 2mm
the diode peak current and average power dissipation so as not to exceed the diode ratings. The main problem with Schottky diodes is that their parasitic capacitance reduces the efficiency, usually negating the possible benefits for LTC3561A circuits. Another problem that a Schottky diode can introduce is higher leakage current at high temperatures, which could reduce the low current efficiency. Remember to keep lead lengths short and observe proper grounding (see Board Layout Considerations) to avoid ringing and increased dissipation when using a catch diode. Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT/VIN. To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT (VIN - VOUT ) VIN
46m 1.8mm 43m 1.5mm 45m 1.8mm 47m 49m 22m 80m 70m 72m 2mm 2mm 3mm 1mm 3mm 3mm
DB3015C-1068AS-1R0N 1.0H DB3018C-1069AS-1R0N 1.0H DB3020C-1070AS-1R0N 1.0H A914BYW-2R2M-D52LC 2.2H A915AY-2ROM-D53LC Coilcraft LPO1704-122ML D01608C-222 LP01704-222M Sumida CR32-1R0 CR5D11-1R0 CDRH3D14-1R2 CDRH4D18C/LD-1R1 CDRH4D28C/LD-1R0 CDRH4D28C-1R1 CDRH4D28-1R2 CDRH6D12-1R0 CDRH4D282R2 CDC5D232R2 Taiyo Yuden NPO3SB1ROM N06DB2R2M N05DB2R2M Murata FDK LQN6C2R2M04 MIPW3226DORGM 2.0H 1.2H 2.2H 2.2H 1.0H 1.0H 1.2H 1.1H 1.0H 1.1H 1.2H 1.0H 2.2H 2.2H 1.0H 2.2H 2.2H 2.2H 0.9H
120m 1mm 40m 1.2mm 36m 1.5mm 24m 22m 2mm 3mm 17.5m 3mm 23.6m 3mm 37.5m 1.5mm 23m 3mm 30m 2.5mm 27m 1.8mm 29m 3.2mm 32m 2.8mm 24m 80m 5mm 1mm
where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - IL/2. This formula has a maximum at VIN = 2VOUT, where IRMS IOUT/2. This simple worst case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling, when not using an all ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required ESR to minimize voltage ripple and load step transients. Typically, once the ESR requirement is satisfied, the capacitance
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Catch Diode Selection Although unnecessary in most applications, a small improvement in efficiency can be obtained in a few applications by including the optional diode D1 shown in Figure 4, which conducts when the synchronous switch is off. In pulse skip mode, the synchronous switch is turned off at a low current and the remaining current will be carried by the optional diode. It is important to adequately specify
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LTC3561A APPLICATIONS INFORMATION
is adequate for filtering. The output ripple (VOUT) is determined by: 1 VOUT IL ESR + 8fO COUT where fO = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. The output ripple is highest at maximum input voltage since IL increases with input voltage. With IL = 0.4 * IOUT(MAX) the output ripple will be less than 100mV at maximum VIN, a minimum COUT value of 10F and fO = 1MHz with: ESRCOUT < 150m Once the ESR requirements for COUT have been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement, except for an all ceramic solution. In surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolytic, special polymer, ceramic and dry tantalum capacitors are all available in surface mount packages. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. Special polymer capacitors, such as Sanyo POSCAP offer very low ESR, , but have a lower capacitance density than other types. Tantalum capacitors have the highest capacitance density, but it has a larger ESR and it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Aluminum electrolytic capacitors have a significantly larger ESR, and is often used in extremely cost-sensitive applications provided that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have the lowest ESR and cost but also have the lowest capacitance density, a high voltage and temperature coefficient and exhibit audible piezoelectric effects. In addition, the high Q of ceramic capacitors along with trace inductance can lead to significant ringing. Other capacitor types include the Panasonic specialty polymer (SP) capacitors. In most cases, 0.1F to 1F of ceramic capacitors should also be placed close to the LTC3561A in parallel with the main capacitors for high frequency decoupling. Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop "zero" at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before their ESR becomes effective. Also, ceramic caps are prone to temperature effects which require the designer to check loop stability over the operating temperature range. To minimize their large temperature and voltage coefficients, only X5R or X7R ceramic capacitors should be used. A good selection of ceramic capacitors is available from Taiyo Yuden, TDK and Murata. Great care must be taken when using only ceramic input and output capacitors. When a ceramic capacitor is used at the input and the power is being supplied through long wires, such as from a wall adapter, a load step at the output can induce ringing at the VIN pin. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, the ringing at the input can be large enough to damage the part. Since the ESR of a ceramic capacitor is so low, the input and output capacitor must instead fulfill a charge storage requirement. During a load step, the output capacitor must instantaneously supply the current to support the load until the feedback loop raises the switch current enough to support the load. The time required for the feedback loop to respond is dependent on the compensation
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LTC3561A APPLICATIONS INFORMATION
components and the output capacitor size. Typically, 3 to 4 cycles are required to respond to a load step, but only in the first cycle does the output drop linearly. The output droop, VDROOP, is usually about 2 to 3 times the linear drop of the first cycle. Thus, a good place to start is with the output capacitor value of approximately: COUT 2.5 IOUT fO * VDROOP Keeping the current small (<5A) in these resistors maximizes efficiency, but making them too small may allow stray capacitance to cause noise problems and reduce the phase margin of the error amp loop. To improve the frequency response, a feed-forward capacitor CF may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Shutdown and Soft-Start The SHDN/RT pin is a dual purpose pin that sets the oscillator frequency and provides a means to shut down the LTC3561A. This pin can be interfaced with control logic in several ways, as shown in Figure 2 and Figure 3. In both configurations, Run = "0" shuts down the LTC3561A and Run = "1" activates the LTC3561A. By activating the LTC3561A, an internal soft-start slowly ramps the output voltage up until regulation. Soft-start prevents surge currents from VIN by gradually ramping the output voltage up during start-up. The output will ramp from zero to full scale over a time period of approximately 0.8ms. This prevents the LTC3561A from having to quickly charge the output capacitor and thus supplying an excessive amount of instantaneous current.
SHDN/RT RT SVIN 1M
More capacitance may be required depending on the duty cycle and load step requirements. In most applications, the input capacitor is merely required to supply high frequency bypassing, since the impedance to the supply is very low. A 10F ceramic capacitor is usually enough for these conditions. Setting the Output Voltage The LTC3561A develops a 0.8V reference voltage between the feedback pin, VFB, and the signal ground as shown in Figure 4. The output voltage is set by a resistive divider according to the following formula: R2 VOUT 0.8V 1+ R1
SHDN/RT RT RUN
3561A F02
RUN
3561A F03
Figure 2. SHDN/RT Pin Activated with a Logic Input
Figure 3. SHDN/RT Pin Activated with a Switch
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LTC3561A APPLICATIONS INFORMATION
Checking Transient Response The OPTI-LOOP(R) compensation allows the transient response to be optimized for a wide range of loads and output capacitors. The availability of the ITH pin not only allows optimization of the control loop behavior but also provides a DC coupled and AC filtered closed loop response test point. The DC step, rise time and settling time at this test point truly reflects the closed loop response. Assuming a predominantly second order system, phase margin and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also be estimated by examining the rise time at the pin. The ITH external components shown in the circuit on page 1 of this data sheet will provide an adequate starting point for most applications. The series R-C filter sets the dominant pole-zero loop compensation. The values can be modified slightly (from 0.5 to 2 times their suggested values) to optimize transient response once the final PC layout is done and the particular output capacitor type and value have been determined. The output capacitors need to be selected because the various types and values determine the loop feedback factor gain and phase. An output current pulse of 20% to 100% of full load current having a rise time of 1s to 10s will produce output voltage and ITH pin waveforms that will give a sense of the overall loop stability without breaking the feedback loop. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD * ESR, where
VIN C6
ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine phase margin. The gain of the loop increases with R and the bandwidth of the loop increases with decreasing C. If R is increased by the same factor that C is decreased, the zero frequency will be kept the same, thereby keeping the phase the same in the most critical frequency range of the feedback loop. In addition, a feedforward capacitor CF can be added to improve the high frequency response, as shown in Figure 4. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Linear Technology Application Note 76. Although a buck regulator is capable of providing the full output current in dropout, it should be noted that as the input voltage VIN drops toward VOUT, the load step capability
OPTI-LOOP is a registered trademark of Linear Technology Corporation.
+
CIN C8 PGND PGND
R6 SVIN PVIN SW SGND ITH LTC3561A VFB R2 RT SGND GND R1
3561A F04
D1 OPTIONAL CF
+
VOUT COUT C5
CITH
RC CC SGND
SGND PGND SHDN/RT
PGND
PGND
SGND SGND
Figure 4. LTC3561A General Schematic
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LTC3561A APPLICATIONS INFORMATION
does decrease due to the decreasing voltage across the inductor. Applications that require large load step capability near dropout should use a different topology such as SEPIC, Zeta or single inductor, positive buck/boost. In some applications, a more severe transient can be caused by switching in loads with large (>1F) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem, if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting.
1 VIN = 3.6V VOUT = 1.2V TO 1.8V fO = 1MHz POWER LOSS (W) 0.1
the losses in LTC3561A circuits: 1) LTC3561A VIN current, 2) switching losses, 3) I2R losses, 4) other losses. 1) The VIN current is the DC supply current given in the electrical characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 2) The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3) I2R Losses are calculated from the DC resistances of the internal switches, RSW, and external inductor, RL. In continuous mode, the average output current flowing through inductor L is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT2(RSW + RL) 4) Other "hidden" losses such as copper trace and internal battery resistances can account for additional efficiency degradations in portable systems. It is very important to include these "system" level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses including diode conduction losses during dead-time and inductor core losses which generally account for less than 2% total additional loss.
Hot Swap is a trademark of Linear Technology Corporation.
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0.01
0.001 0.1
1
10 100 1000 LOAD CURRENT (mA)
10000
3561A F01
Figure 5. Power Loss vs Load Currrent
Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: %Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of
15
LTC3561A APPLICATIONS INFORMATION
Thermal Considerations In a majority of applications, the LTC3561A does not dissipate much heat due to its high efficiency. However, in applications where the LTC3561A is running at high ambient temperature with low supply voltage and high duty cycles, such as in dropout, the heat dissipated may exceed the maximum junction temperature of the part. If the junction temperature reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. To avoid the LTC3561A from exceeding the maximum junction temperature, the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated exceeds the maximum junction temperature of the part. The temperature rise is given by: TRISE = PD * JA where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TRISE + TAMBIENT As an example, consider the case when the LTC3561A is in dropout at an input voltage of 3.3V with a load current of 1A. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) resistance of the P-channel switch is 0.17. Therefore, power dissipated by the part is: PD = I2 * RDS(ON) = 170mW The DD8 package junction-to-ambient thermal resistance, JA, will be in the range of about 43C/W. Therefore, the junction temperature of the regulator operating in a 70C ambient temperature is approximately: TJ = 0.17 * 43 + 70 = 77.31C Remembering that the above junction temperature is obtained from an RDS(ON) at 25C, we might recalculate the junction temperature based on a higher RDS(ON) since it increases with temperature. However, we can safely assume that the actual junction temperature will not exceed the absolute maximum junction temperature of 125C. Design Example As a design example, consider using the LTC3561A in a portable application with a Li-Ion battery. The battery provides a VIN = 2.5V to 4.2V. The load requirement is a maximum of 1A, but most of the time it will be in standby mode, requiring only 10mA. The output voltage is VOUT = 1.8V. Since the load still needs power in standby, Burst Mode operation is selected for good low load efficiency. First, calculate the timing resistor for 1MHz operation: RT = 5 * 107 (103)-1.6508 = 557.9k Use a standard value of 549k. Next, calculate the inductor value for about 40% ripple current at maximum VIN: L= 1.8V 1.8V * 1- = 2.57H 1MHz * 400mA 4.2V
Choosing the closest inductor from a vendor of 2.2H, results in a maximum ripple current of: IL = 1.8V 1.8V * 1- = 468mA 1MHz * 2.2H 4.2V
For cost reasons, a ceramic capacitor will be used. COUT selection is then based on load step droop instead of ESR requirements. For a 5% output droop: COUT 2.5 1A 27F 1MHz * (5% * 1.8V)
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LTC3561A APPLICATIONS INFORMATION
The closest standard value is 22F Since the output . impedance of a Li-Ion battery is very low, CIN is typically 10F In noisy environments, decoupling SVIN from PVIN . with an R6/C8 filter of 1/0.1F may help, but is typically not needed. For the feedback resistors, choose R1 = 200k, R2 can be calculated from: V 1.8 V - 1 * 200k = 250k R2 = OUT - 1 * R1 = 0.8 V 0.8 Choose a standard value of 249k for R2. The compensation should be optimized for these components by examining the load step response but a good place to start for the LTC3561A is with a 16.9k and 680pF filter. The output capacitor may need to be increased depending on the actual undershoot during a load step. Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3561A. These items are also illustrated graphically in the layout diagram of Figure 6. Check the following in your layout: 1. Does the capacitor CIN connect to the power VIN (Pin 5) and power GND (Pin 4) as close as possible? This capacitor provides the AC current to the internal power MOSFETs and their drivers. 2. Are the COUT and L1 closely connected? The (-) plate of COUT returns current to PGND and the (-) plate of CIN. 3. The resistor divider, R1 and R2, must be connected between the (+) plate of COUT and a ground line terminated near SGND (Exposed Pad). The feedback signal VFB should be routed away from noisy components and traces, such as the SW line (Pin 3), and its trace should be minimized. 4. Keep sensitive components away from the SW pin. The input capacitor CIN, the compensation capacitor CC and CITH and all the resistors R1, R2, RT, and RC should be routed away from the SW trace and the inductor L1. The SW pin pad should be kept as small as possible. 5. A ground plane is preferred, but if not available, route all small-signal components back to the SGND pin (Exposed Pad). All SGND and PGND pins must be connected together through a thick copper trace or ground plane. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to the Exposed Pad for best results.
CIN VIN PVIN PGND COUT L1 VOUT
SVIN SW LTC3561A VFB SGND ITH R2 C4 R1 RC CC CITH SHDN/RT RT
3561A F06
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 6. LTC3561A Layout Diagram (See Board Layout Checklist)
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LTC3561A TYPICAL APPLICATION
General Purpose Buck Regulator Using Ceramic Capacitors
VIN 2.5V TO 5.5V C1 22F PGND SVIN PVIN LTC3561A SW R2 249k VFB ITH SGND R3 16.9k C3 680pF SGND GND SHDN/RT 1.8V PGND R4 549k R1A 200k R1B 287k 1.5V 1.2V C4 22pF R1C 499k C2 22F L1 2.2H VOUT 1.2V/1.5V/1.8V AT 1A
3561A TA02a
SGND
PGND
NOTE: IN DROPOUT, THE OUTPUT TRACKS THE INPUT VOLTAGE C1, C2: TAIYO YUDEN JMK325BJ226MM L1: TOKO A914BYW-2R2M (D52LC SERIES)
Efficiency vs Output Current
100 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 100 1000 OUTPUT CURRENT (mA) 10000
3561A TA02b
VOUT = 1.2V
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.2V ILOAD = 20mA TO 1A
3561A TA02c
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV VIN = 3.6V 40s/DIV VOUT = 1.2V ILOAD = 200mA TO 1A
3561A TA02d
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LTC3561A PACKAGE DESCRIPTION
DD Package 8-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1698)
0.675 0.05
3.5 0.05 1.65 0.05 2.15 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS R = 0.115 TYP 5
0.38 0.10 8
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (NOTE 6)
1.65 0.10 (2 SIDES)
(DD) DFN 1203
0.200 REF
0.75 0.05
4 0.25 0.05 2.38 0.10 (2 SIDES)
1 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3561A TYPICAL APPLICATIONS
1mm Height, 2MHz, Li-Ion to 1.8V Converter
VIN 2.5V TO 4.2V PVIN C1 10F SVIN LTC3561A VFB SHDN/RT R4 178k R2 R1 200k 249k C1, C2: TAIYO YUDEN JMK107BJ106MA L1: FDK MIPW3226DORGM SW L1 0.9H C4 22pF C2 10F 2 VOUT 1.8V AT 1A
ITH R3 16.9k SGND PGND C3 470pF
3561A TA04a
Efficiency vs Output Current
100 90 80 70 EFFICIENCY (%) 60 50 40 30 20 10 0 1 VIN = 2.7V VIN = 3.6V VIN = 4.2V 10 100 1000 OUTPUT CURRENT (mA) 10000
3561A TA04b
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 30mA TO 1A
3561A TA04c
VOUT 100mV/DIV AC COUPLED IL 1A/DIV ILOAD 1A/DIV 40s/DIV VIN = 3.6V VOUT = 1.8V ILOAD = 200mA TO 1A
3561A TA04d
RELATED PARTS
PART NUMBER LTC3406/LTC3406B LTC3407/LTC3407B LTC3410/LTC3410B LTC3411A LTC3412A LTC3531/LTC3531-3 LTC3531-3.3 LTC3532 LTC3542 LTC3544 LTC3547/LTC3547B LTC3548/LTC3548-1 LTC3548-2 LTC3560 DESCRIPTION 600mA (IOUT), 1.5MHz Synchronous Step-Down DC/DC Converters Dual 600mA/800mA (IOUT), 1.5MHz/2.25MHz Synchronous Step-Down DC/DC Converters 300mA (IOUT), 2.25MHz Synchronous Step-Down DC/DC Converters 1.25A (IOUT), 4MHz Synchronous Step-Down DC/DC Converter 2.5A (IOUT), 4MHz Synchronous Step-Down DC/DC Converter 200mA (IOUT), 1.5MHz Synchronous Buck-Boost DC/DC Converters 500mA (IOUT), 2MHz Synchronous Buck-Boost DC/DC Converter 500mA (IOUT), 2.25MHz Synchronous Step-Down DC/DC Converter Quad 300mA + 2x 200mA + 100mA, 2.25MHz Synchronous Step-Down DC/DC Converter Dual 300mA, 2.25MHz Synchronous Step-Down DC/DC Converters Dual 400mA/800mA, (IOUT), 2.25MHz Synchronous Step-Down DC/DC Converters 800mA (IOUT), 2.25MHz Synchronous Step-Down DC/DC Converter COMMENTS 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20A, ISD < 1A, ThinSOTTM 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD < 1A, MS10E, DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26A, ISD < 1A, SC70 96% Efficiency, VIN: 2.6V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60A, ISD < 1A, MS10, 3mm x 3mm DFN 96% Efficiency, VIN: 2.6V to 5.5V, VOUT(MIN) = 0.8V, IQ = 62A, ISD < 1A, TSSOP16E, 4mm x 4mm QFN 95% Efficiency, VIN: 1.8V to 5.5V, VOUT(MIN) = 2V to 5V, IQ = 16A, ISD < 1A, ThinSOT, DFN 95% Efficiency, VIN: 2.4V to 5.5V, VOUT(MIN) = 2.4V to 5.25V, IQ = 35A, ISD < 1A, MS10, DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 26A, ISD < 1A, 2mm x 2mm DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 70A, ISD < 1A, 3mm x 3mm QFN 96% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD < 1A, 2mm x 3mm DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD < 1A, MS10E, DFN 95% Efficiency, VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 16A, ISD < 1A, ThinSOT
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ThinSOT is a trademark of Linear Technology Corporation.
20 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
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